Multicarrier communication systems, including the most commonly used DMT and OFDM communication systems, have attracted considerable attention in a variety of high-speed communication applications, including digital subscriber line (DSL), digital terrestrial broadcasting, wireless local area network (WLAN), wireless metropolitan area network (WMAN), dedicated short range communication (DSRC), power line communication, and so on. They also show promise as future generation of mobile communication systems. The advantage of multicarrier communication systems comes from dividing high-speed data stream into multiple parallel portions of data streams transmitted via individual sub-carriers. Each portion of data stream is transmitted at a lower speed and thus robust against the effects of channel impairments such as multipath fading and impulse noise.
FIG. 1 is a simplified block diagram illustrating a typical OFDM transmitter. As can be seen from the OFDM transmitter, data X[k], k=0, 1, . . . , N−1, to be transmitted within an OFDM symbol period, are transformed via a serial/parallel (S/P) converter 10, an N-point inverse fast Fourier transform (N-IFFT) 20, and a parallel/serial (P/S) converter 30 into the following baseband transmitted signal:
                                                                                          x                  ⁡                                      [                    n                    ]                                                  =                                                      1                                          N                                                        ⁢                                                            ∑                                              k                        =                        0                                                                    N                        -                        1                                                              ⁢                                                                  X                        ⁡                                                  [                          k                          ]                                                                    ⁢                                              W                        N                        kn                                                                                                        ,                                                                          n                =                0                            ,              1              ,              …              ⁢                                                          ,                              N                -                1                                                                        (        1        )            whereWN≡ej2π/N  (2)is the twiddle factor. The discrete-time transmitted signal x[n] given by (1) is then inserted with cyclic prefix followed by digital-to-analog (D/A) conversion, and the resultant analog signal x(t) is sent to RF front end for further processing including in-phase/quadrature-phase (I/Q) modulation, up conversion and power amplification.
It is known that the PAPR of the analog signal x(t) is higher than that of the discrete-time counterpart x[n] by several dB, and can be approximated by that of x[n/R] where x[n/R] denotes the signal taken from R times oversampling of x[n]. Note that the cyclic prefix insertion 40 has no any effect on the PAPR level of x(t). Thus, it is convenient to evaluate the PAPR level of x(t) in terms of that of x[n/R] given by
                    PAPR        =                                            max                              0                ≤                n                ≤                                  RN                  -                  1                                                      ⁢                                                                            x                  ⁡                                      [                                          n                      /                      R                                        ]                                                                              2                                            E            ⁢                          {                                                                                      x                    ⁡                                          [                                              n                        /                        R                                            ]                                                                                        2                            }                                                          (        3        )            where E{ } denotes expectation operation. Typically, the approximation is quite accurate for R≧4.
One major drawback of multicarrier communication systems is the high PAPR of the baseband transmitted signal x(t). When passing through an RF front end without sufficient power back-offs, the signal x(t) will be distorted by the nonlinearity of RF power amplifier. In particular, the nonlinearity will incur not only the in-band signal distortion leading to bit-error-rate (BER) performance degradation, but also the out-of-band radiation (or spectrum re-growth) leading to adjacent channel interference and violation of government's spectrum regulation.
Conventional solution to this problem is simply utilizing a power amplifier with large linear range and large power back-offs at the expense of low power efficiency, high power consumption, and high manufacturing cost. Alternatively, the problem can be resolved by using PAPR reduction approaches. One of the PAPR reduction approaches is block-coding approach, which tries to find out a coding rule so that all the encoded codewords result in very low PAPR levels for the transmitted signal x(t). However, extremely low code rate and extraordinary encoding/decoding complexity make the approach only suitable for systems with small constellation size and small number of sub-carriers.
Another PAPR reduction approach is deliberately clipping approach. In this approach, those amplitude levels of the transmitted signal exceeding a certain threshold are clipped, and the clipped signal is filtered to eliminate out-of-band radiation. Nevertheless, large clipping distortion may lead to severe BER performance degradation and inadequately filtering may lead to peak re-growth. On the other hand, probabilistic approach tries to reduce the probability of high PAPR level for the transmitted signal by changing the phase, order, level or other properties of the data stream. Probabilistic approach includes partial transmit sequence (PTS) method, selective mapping (SLM) method, tone reservation (TR) method, tone injection (TI) method, and pulse superposition method, among which the PTS method seems to be most attractive in terms of implementation complexity as well as PAPR reduction performance.
For a number of PAPR reduction methods such as the PTS, SLM and pulse superposition methods, the associated receiver needs to know about the modifications (e.g., the modified phases, orders or levels) that have been made to the data stream at the transmitter during PAPR reduction procedure. The modifications are referred to as the side information, which is used for correctly recovering the original data stream at the receiver. Correspondingly, the reliability of transmitting the side information related to PAPR reduction is extremely important for system's functionality.
FIG. 2 is the block diagram of an OFDM transmitter using the PTS method, which is disclosed in the U.S. Pat. No. 6,125,103. For reducing the PAPR of x(t), the input data block X=[X[0], X[1], . . . , X[N−1]]T is first partitioned into L disjoint sub-blocks (or clusters), denoted by X1, X2, . . . , XL, of length N where the superscript ‘T’ represents vector transposition. Only NIL entries of Xl, lε{1, 2, . . . , L}, are taken from the corresponding entries of X and the remaining ones are set to zero. The partition scheme can be interleaved, adjacent, or irregular. The L disjoint sub-blocks are then phase-rotated and combined to form the following signal:
                              X          ~                =                              ∑                          l              =              1                        L                    ⁢                                    b              l                        ⁢                          X              l                                                          (        4        )            or, equivalently,
                                                                                                              X                    ~                                    ⁡                                      [                    k                    ]                                                  =                                                      ∑                                          l                      =                      1                                        L                                    ⁢                                                            b                      l                                        ⁢                                                                  X                        l                                            ⁡                                              [                        k                        ]                                                                                                        ,                                                                          k                =                0                            ,              1              ,              …              ⁢                                                          ,                              N                -                1                                                                        (        5        )            where bl is the phase rotation factor (i.e., |bl|=1) associated with the lth sub-block Xl.
Taking N-IFFT of (5) yields the transmitted signal
                                                                                                              x                    ~                                    ⁡                                      [                    n                    ]                                                  =                                                      ∑                                          l                      =                      1                                        L                                    ⁢                                                            b                      l                                        ⁢                                                                  x                        l                                            ⁡                                              [                        n                        ]                                                                                                        ,                                                                          n                =                0                            ,              1              ,              …              ⁢                                                          ,                              N                -                1                                                                        (        6        )            where x[n], representing the N-IFFT of Xl[k], is referred to as the PTS. The goal of the PTS method is to search for optimal phase sequence {b1, b2, . . . , bL} such that the PAPR level of the resultant transmitted signal is minimum. In practice, the phase of bl is admitted to be one of a limited set of discrete values {2πm/M, m=0, 1, . . . , M−1}, and b1 can be fixed to unity without sacrificing any PAPR reduction performance. As such, finding optimal phase sequence {b2, b3, . . . , bL} requires performing M(L−1) computations of (3), implying that optimal search for {b2, b3, . . . , bL} is almost prohibitive for large L and M. For this reason, there have been low-complexity sub-optimal search algorithms, for which {b2, b3, . . . , bL} is selected from a smaller subset of all possible candidates of {b2, b3, . . . , bL}. Obviously, sub-optimal search algorithms suffer from some degradation in PAPR reduction performance.
The phase sequence {b2, b3, . . . , bL} is considered as the side information to be transmitted to the associated receiver, so that the receiver can correctly recover the data stream of sub-blocks Xl, l=2, 3, . . . , L. In conventional PTS method, the side information is transmitted via (L−1) reserved sub-carriers where one sub-carrier within each sub-block Xl(lε{2, 3, . . . , L}) is allocated. The conventional method, however, provides no protection capability for the side information over these reserved sub-carriers and, thus, may result in unreliable side information detection at the receiver under noisy channel conditions. On the other hand, our invention provides a low-complexity and low-latency method and apparatus for reliably transmitting the side information regarding PAPR reduction in multicarrier communication systems.